Low voltage operational amplifier and method

ABSTRACT

Low voltage operational amplifier (10) operates in a voltage range of one to eight volts over a temperature range of 0° to 70° centigrade. Op amp input stage (12) uses N-channel depletion-mode MOSFETs to provide amplification of the differential input and maintain constant transconductance. Source follower MOSFET (13) provides unity gain in transferring the AC signal, STAGE-1 OUTPUT, to the base of current sinking transistor (18). Sink control circuit (14) and source control circuit (22) generate the base drive currents for transistors (18) and (24). The signal at the output of MOSFET (13) either causes the sink transistor (18) to sink current or the signal to be transposed by means of a translinear loop (16) and causes the source transistor (24) to source current.

BACKGROUND OF THE INVENTION

The present invention relates in general to integrated circuit designand, more particularly, to monolithic operational amplifiers having adifferential amplifier input stage that employs depletion mode MetalOxide Semiconductor Field Effect Transistor (MOSFET) devices forattaining rail-to-rail input capability.

The industry trend for electronic systems which encompass operationalamplifiers is toward lower operating voltages supplied from batterysources. Thus, amplifiers are used in applications requiring low voltagesingle supply operations in addition to traditional op amp applicationsuch as high input impedance, low input offset voltage, low noise, highbandwidth, high speed and sufficient output drive capabilities.Different manufacturing processes for integrated circuits have allowedtechniques for differential input stages such as darlington PNPtransistors and P-channel depletion-mode MOSFETS, aimed at satisfyingthe mentioned criteria for input stages of op amps. Amplifier outputstages have used techniques involving combinations of transistors thatinclude NPN, PNP and MOSFETs, aimed at low crossover distortion, largeoutput voltage swings including rail to rail performance, excellentphase and gain margins, low output impedance and symmetrical source andsink capabilities.

Although the various types of input stages operate from a single supplyvoltage source, the low voltage limit for amplifier operation differsfor each type of input stage and each integrated circuit manufacturingprocess. Present input stage designs for op amps exhibit voltageoperation limits that hinder applications in products powered bybatteries having an end of life near one volt. For example, an op ampusing multiple bipolar transistors for compensating temperature effectsand current paths have low operating voltage limitations imposed bystandard transistor base to emitter voltage drops.

Hence, a need exists for a versatile operational amplifier that can beused in a variety of applications powered from battery sources,especially low voltage applications that do not diminish thecharacteristics of an operational amplifier. A need exists for an op ampinput stage that provides high input impedance and a low input offsetvoltage. A need exists for an op amp that minimizes transistors in thesignal path for providing high speed and high bandwidth and still haveboth input and output rail to rail capabilities.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an operational amplifier, in accordancewith a preferred embodiment of the present invention;

FIG. 2 is a schematic diagram showing a preferred embodiment of an inputstage for the low voltage operational amplifier shown in FIG. 1;

FIG. 3 is a schematic diagram showing an alternate embodiment of aninput stage for the low voltage operational amplifier shown in FIG. 1;

FIG. 4 is a schematic diagram showing another alternate embodiment of aninput stage for the low voltage operational amplifier shown in FIG. 1;

FIG. 5 is a schematic diagram showing an output sink transistor basecurrent generating stage for the operational amplifier shown in FIG. 1;

FIG. 6 is a schematic diagram showing an output source transistor basecurrent generating stage for the operational amplifier shown in FIG. 1;

FIG. 7 is a schematic diagram showing an alternate embodiment of a lowvoltage translinear loop for the operational amplifier shown in FIG. 1;and

FIG. 8 is a schematic diagram showing a preferred embodiment of a lowvoltage translinear loop for selecting the source or sink capabilitiesof an output amplifier as shown in FIG. 1.

DETAILED DESCRIPTION OF THE DRAWINGS

The block diagram for low voltage operational amplifier 10 is shown inFIG. 1. The differential input signal V_(IN) is applied across the twoinputs to op amp input stage 12. Terminal 67 of op amp input stage 12 iscoupled to the gate of MOSFET 13. A MOSFET device with a drain terminal,source terminal, and gate terminal is a current conducting transistorwith a first current terminal, a second current terminal, and a controlterminal. Note that MOSFETs or other equivalents can be used whereappropriate instead of bipolar transistors in the followingdescriptions. The drain of MOSFET 13 is coupled to power supplyconductor V_(CC) operating at a positive power supply, such as one volt.The negative supply for operational amplifier 10 is shown in the figuresand described throughout as ground reference. The source of MOSFET 13 iscoupled to the input of sink control circuit 14 and to the firstterminal of current sink 15, sinking approximately twenty-fivemicroamps. The bulk of MOSFET 13 (not shown) is coupled to a voltagereference (not shown). The second terminal of current sink 15 is coupledto ground reference. Terminal 107 of sink control circuit 14 is coupledto a first input of translinear loop 16 and to the base of NPNtransistor 18. Capacitor 20 is coupled between the base and collector oftransistor 18 and in the preferred embodiment has a capacitance ofapproximately eight picofarads. An NPN transistor or a PNP transistorwith an emitter terminal, collector terminal, and base terminal is acurrent conducting transistor with a first current terminal, a secondcurrent terminal, and a control terminal. The emitter of transistor 18is coupled to ground reference while the collector of transistor 18 iscoupled to terminal 25 for providing the output signal, V_(OUT).

Terminal 147 of source control circuit 22 in FIG. 1 is coupled to theoutput of translinear loop 16 and to the base of PNP transistor 24.Capacitor 26 is coupled between the base and collector of transistor 24and in the preferred embodiment has a capacitance of approximately eightpicofarads. The emitter of transistor 24 is coupled to operatingpotential V_(CC). The collector of transistor 24 is coupled to terminal25 for providing signal V_(OUT) as the output driver stage output.Capacitor 28, selected at approximately twenty picofarads, and resistor27, selected at about 1.4 kilohms, are serially coupled between terminal25 and terminal 67 of op amp input stage 12.

Low voltage operational amplifier 10 has two stages of amplification.The output of op amp input stage 12 comprises an amplified differentialinput signal as the first stage of amplification and output driver stage29 provides the second stage of amplification. MOSFET 13 is connected asan N-channel depletion-mode source follower MOSFET and processed to havea negative threshold voltage. In a depletion-mode source follower, thevoltage potential imposed at the gate terminal is passed to the sourceterminal. The MOSFET device does not change or amplify the input signaland is therefore providing unity gain in transferring the receivedoutput from op amp input stage 12. MOSFET 13 provides a high inputimpedance inherent in MOSFET devices. The high input impedance is basedon the isolation of the gate terminal from current paths, either toground reference or to operating potential V_(CC), due to dielectricoxides formed in processing MOSFET devices.

Referring to FIG. 1, sink control circuit 14 generates the base currentdrive for transistor 18 that controls low voltage operational amplifier10 current sinking capabilities. Low voltage operational amplifier 10has a V_(CC) operating range of eight volts to one volt. At an operatingpotential V_(CC) of three volts, the current sinking capability oftransistor 18 is fifty milliamps. Source control circuit 22 generatesthe base current drive for transistor 24 that controls low voltageoperational amplifier 10 current sourcing capabilities. At an operatingpotential V_(CC) of three volts, the current sourcing capability oftransistor 24 is fifty milliamps. When the signal V_(IN) is amplified byop amp input stage 12, the signal to translinear loop 16 at terminal 107is the transferred output of the signal at terminal 67. Thus, based onthe input signal V_(IN) to op amp input stage 12, translinear loop 16selects whether sink control circuit 14 is operative and low voltageoperational amplifier 10 is sinking current through transistor 18 orsource control circuit 22 is operative and low voltage operationalamplifier 10 is sourcing current through transistor 24.

Referring to FIG. 1, low voltage operational amplifier 10, with twostages of amplification, has two frequency poles. The function ofresistor 27 and capacitor 28 is to move one frequency pole higher thanthe bandwidth of low voltage operational amplifier 10 and cause theother dominant frequency pole to move lower in frequency. The purpose ofthis pole splitting technique is to ensure amplifier stability. That is,by moving the second pole out beyond the unity gain point sufficientphase margin is achieved such that the phase shift is not one hundredand eighty degrees at the unity gain point, and low voltage operationalamplifier 10 is prevented from oscillating.

FIG. 2 shows a schematic of a preferred embodiment of op amp input stage12 suitable for use with the op amp shown in FIG. 1. The first stage ofamplification in low voltage operational amplifier 10 is accomplished byop amp input stage 12. Signal V_(IN) is the differential input coupledacross the gates of N-channel depletion-mode Metal Oxide SemiconductorField Effect Transistors (MOSFETs) 30 and 32. The drain of MOSFET 30 iscoupled to one terminal of current source 34, supplying approximatelyeighty microamps of current. The drain of MOSFET 32 is coupled to oneterminal of current source 36, supplying approximately eighty microampsof current. The second terminals for both current sources 34 and 36 arecoupled to operating potential V_(CC). Both source terminals of MOSFETS30 and 32 are coupled to one terminal of current sink 38, sinkingapproximately forty microamps. The other terminal of current sink 38 iscoupled to ground reference. The bulk, or well, terminals for bothMOSFET 30 and MOSFET 32 are coupled to ground reference.

The differential pair of MOSFETS 30 and 32 in FIG. 2 that receive theinput signal V_(IN) provide two outputs from the drain terminals ofMOSFETS 30 and 32, supplied as Alternating Current (AC) signal inputs tocurrent bias circuit 39. The function of current bias circuit 39 is toprovide equal loads on the two inputs coupled from the drain terminalsof MOSFETS 30 and 32, match source and sink current capabilities atoutput terminal 67, provide a high impedance at output terminal 67, andperform a differential to single ended conversion of the input signalV_(IN). Transistors 40, 42, 44, 46, and 48 are PNP type with commontransistor base terminals coupled to the collector of transistor 48 in apreferred embodiment. Current sink 50, sinking approximately twentymicroamps, has a first terminal coupled to the common base and collectorterminals of transistor 48. The second terminal of current sink 50 iscoupled to ground reference. The emitters of transistors 40 and 42 arecoupled to the drain of MOSFET 30. The emitters of transistors 44 and 46are coupled to the drain of MOSFET 32. The emitter of transistor 48 iscoupled to one terminal of resistor 49, selected at approximately 7.5kilohms, and the second terminal of resistor 49 is coupled to operatingpotential V_(CC).

Transistors 52, 54, 56, 58, 60, 62, 64, 66, and 72 are NPN type in apreferred embodiment of op amp input stage 12 in FIG. 2. The commoncollectors of transistors 44 and 52 couple to the common bases oftransistors 54 and 56. Common collectors of transistors 40, 42, 58, and60, are coupled to the common bases of transistors 62 and 64. Theemitter of transistor 52 is coupled to the collector of transistor 54.The collector of transistor 56 is coupled to the emitter of transistor58. Emitters of transistors 54 and 56 are coupled to ground reference.The emitter of transistor 60 is coupled to the collector of transistor62. The collector of transistor 64 is coupled to the emitter oftransistor 66. The emitters of transistors 62 and 64 are coupled toground reference. The common base terminals of transistors 52, 58, 60,and 66 are coupled to one terminal of current source 68, which sourcestwenty microamps, and to one terminal of a nine kilohm resistor 70. Thesecond terminal of current source 68 is coupled to operating potentialV_(CC). The second terminal of resistor 70 is coupled to the commoncollector and base of transistor 72. The emitter of transistor 72 iscoupled to ground reference. The common collectors of transistors 46 and66 are coupled to output terminal 67 for providing signal STAGE-1 OUTPUTas the op amp input stage output. This completes the connections for opamp input stage 12.

As one feature of the present invention, op amp input stage 12 usesN-channel depletion-mode MOSFETS 30 and 32 to swing rail to rail andexhibit minimal transconductance changes, whether gates are at ground,operating supply or half supply. Transconductance is measured as thechange in MOSFET drain current for a given change in MOSFETgate-to-source voltage. Bandwidth of the amplifier is proportional tothe transconductance. MOSFET 13 found in FIG. 1 and MOSFETS 30 and 32 ofop amp input stage 12 are N-channel depletion-mode devices built on asilicon substrate having four terminals represented as gate, drain,source, and bulk. A processing mask layer defines the region forimplanting N-type doping material, such as arsenic, into the silicon toform source and drain regions. The MOSFET gate region is also defined bya processing mask layer such that the gate conductor and gate oxidephysically separate the source and drain regions. N-channel source anddrain regions are confined within a well region for receiving a p-typematerial implant, such as boron. Low resistance conducting materials,such as aluminum metal, provide electrical connections to the gateterminal, source terminal, drain terminal, and the well terminal, orbulk.

Op amp input stage 12 in FIG. 2 accepts small signal differential inputsand accurately provides amplification. N-channel depletion-mode MOSFETS30 and 32 continually operate in the saturation mode over the voltagerange of input signal V_(IN) and over the range of operating potentialV_(CC). Since MOSFET devices operate in the saturation region when thedevice drain voltage is greater than the difference of device gatevoltage and threshold voltage, device threshold voltage becomes animportant MOSFET parameter. For depletion-mode MOSFETS 13, 30 and 32threshold voltage is a measured gate to source voltage at whichdrain-to-source current conduction is terminated.

The threshold voltage for an N-channel device fabricated on a siliconwafer is defined as the gate voltage required to overcome fourparticular physical processing fabrication effects to eliminate thedrain-to-source conduction channel and terminate current flow. The firstand second threshold effects are based on the flat-band voltage, definedas the voltage potential applied at the gate to overcome the workfunction and the charges under the gate at the silicon-silicon dioxideinterface. Work function potential is based on the difference ofelectron energies at the Fermi level in the gate material and in thesemiconductor material. Charges at the silicon-silicon dioxide interfaceare dependent on crystalline orientation and integrated circuitprocessing. The third and fourth threshold voltage effects for a MOSFETare attributed to the voltage potential required to form a surfaceinversion layer. The N-type conduction channel layer induced from sourceto drain by an electric field applied at the gate conductor depends onthe concentration of impurities in the bulk material.

The threshold voltage term for an N-channel depletion-mode MOSFET isbased on four terms directly related to processing during themanufacture of integrated circuits, such as wafer starting material,type of conducting gate material, impurities in the silicon at the gateoxide interface, and doping concentrations of the P-well bulk region. Aprocessing flow step, known as threshold adjust implant, allows theN-channel MOSFET device to be altered from enhancement-mode todepletion-mode by imposing heavier N-type dose implants in the gateregion. Depletion-mode MOSFETS 30 and 32 are processed with a negativethreshold voltage. Even with the gate at ground reference, a MOSFETdepletion-mode device with a negative threshold value has established aninversion layer for a current conduction path from the drain-to-sourceterminals.

With the gates of depletion-mode MOSFETS 30 or 32 at ground reference,the devices are saturated and operating in the normal common mode rangewith minimal body effect. N-channel MOSFETs are desirable because of thehigh transconductance per device area as processed on a silicon wafer.As the gate voltage potential of MOSFETS 30 and 32 rise above groundreference, the source terminals of MOSFETS 30 and 32 follow the gatevoltage positive. With the bulk terminals of MOSFETS 30 and 32 coupledto ground reference, source terminal voltage potentials above the bulkterminal voltages cause the channel conductance to be modulated, whichis body effect. An increase in source to bulk voltage dynamically shiftsthe threshold voltage of N-channel depletion-mode MOSFET device from anegative value, induced by implant doping in the bulk, toward a positivevalue. With a positive threshold value, the common mode range of theMOSFET device shifts toward sensing at the positive supply rail. HeavyP-type well doping increases the body effect of N-channel depletion-modeMOSFETS 30 and 32 to maintain operation of both devices in thesaturation region while operating at the positive rail. Therefore, bodyeffect aids the N-channel depletion-mode MOSFET device by modulating thethreshold voltage and keeping the MOSFET device operating in thesaturation region.

An alternate embodiment of op amp input stage 12 involves replacing thefour transistors 52, 54, 56, and 58 by two NPN transistors arranged as acurrent mirror and replacing the four transistors 60, 62, 64, and 66 bytwo NPN transistors also arranged as a current mirror. In referring toFIG. 2, the alternate embodiment in effect places a wire short fromcollector to emitter for each of transistors 52, 58, 60, and 66 and thenremoving those transistors from the schematic. In the alternateembodiment, the voltage reference provided by current source 68,resistor 70, and transistor 72 of op amp input stage 12 is removed.

Referring to op amp input stage 12 shown in FIG. 2 with the alternateembodiment just described, the current flowing in the collector oftransistor 40 is I_(ce), approximately thirty microamps. An equivalentcurrent I_(ce) also flows in each of transistors 42, 44, and 46 when theop amp inputs are in common mode. A 2I_(be) portion of the I_(ce)collector current in transistor 44 is used to supply base currents totransistors 54 and 56, leaving (I_(ce) -2I_(be)) current in thecollector of transistor 54. The current mirror of transistors 54 and 56implies that an (I_(ce) -2I_(be)) current is also in the collector oftransistor 56. With transistors 40 and 42 each supplying equal I_(ce)currents, and current in the collector of transistor 56 at (I_(ce)-2I_(be)), the collector current in transistor 62 is I_(ce) aftersubtracting the current 2I_(be) to the bases of transistors 62 and 64.The current mirror of transistors 62 and 64 implies that an equivalentI_(ce) collector current of transistor 62 is collector current intransistor 64, respectively matching the I_(ce) current supplied bytransistor 46. Thus, source and sink current to the STAGE-1 OUTPUTsignal have matching output capabilities through sourcing current intransistor 46 and sinking current in transistor 64.

The alternate embodiment in the simplified form just described wasenhanced to the preferred embodiment shown in FIG. 2 for the purpose ofimproving the effective output impedance for the signal STAGE-1 OUTPUTat output terminal 67. The addition of cascode transistor 66 in serieswith transistor 64 increases the output impedance at output terminal 67.Transistor 60 is added to balance transistor 66. The addition oftransistors 52 and 58 to transistors 54 and 56 form another cascodecurrent mirror for matching and canceling I_(be) currents to the cascodecurrent mirror formed by transistors 60, 62, 64 and 66.

Op amp input stage 12 as shown in FIG. 2 provides the first stage ofsignal V_(IN) amplification based on the saturation currents of MOSFETS30 and 32 obeying a square law relationship to the voltage applied atthe gate terminal. Current bias circuit 39, with terminal 67 supplyingthe STAGE-1 OUTPUT signal, is a high impedance output consideringconnection to the common collectors of transistors 46 and 66. Currentbias circuit 39 also matches source and sink current capabilities oftransistors 46 and 66 in supplying the STAGE-1 OUTPUT signal. Asdescribed above, transistors 52, 54, 56, and 58 are coupled together ina manner allowing I_(be) cancellation such that transistors 46 and 66match source and sink current capabilities in supplying the STAGE-1OUTPUT signal at terminal 67.

Referring to FIG. 2, the base coupled to the collector of transistor 48sets a V_(be) diode voltage reference and, when added to the approximatetwenty microamps of current through resistor 49 from current sink 50,sets a voltage approximately 0.75 volts below operating potentialV_(CC). Transistors 40, 42, 44, and 46 are kept in the active operatingregion by the 0.75 volts supplied as the transistor base referencevoltage below operating potential V_(CC). Likewise, an approximate 0.75volt potential above ground reference voltage is used to biastransistors 52, 58, 60, and 66 in their active regions. The 0.75 voltpotential is the combination of twenty microamps of current from currentsource 68, through nine kilohm resistor 70 in addition to the V_(be)voltage drop of transistor 72.

FIG. 3 shows another alternate embodiment of op amp input stage 12.MOSFETs 30 and 32 are coupled to current sources 34 and 36 and tocurrent sink 38 as discussed above. The differential pair of MOSFETS 30and 32 that receive the input signal V_(IN) provide two outputs fromdrain terminals of MOSFETS 30 and 32. The output from the drain ofMOSFET 30 is coupled to the emitter of PNP transistor 200. The outputfrom the drain of MOSFET 32 is coupled to the emitter of PNP transistor202. Common bases of transistors 200 and 202 are coupled to receive avoltage reference. The common bases of NPN transistors 204 and 206 arecoupled to the collector of transistor 204. The collector of transistor200 is coupled to the collector of transistor 204. The collector oftransistor 202 is coupled to terminal 67 for providing the output signalSTAGE-1 OUTPUT. The collector of transistor 206 is coupled to terminal67. The emitters of transistors 204 and 206 are coupled to groundreference.

Again referring to FIG. 3, the differential pair of MOSFETs 30 and 32receive the input signal V_(IN) and along with transistors 200, 202,204, and 206 perform a differential to single ended conversion of theinput signal. However, transistors 202 and 206 do not match source andsink current capabilities or provide as high an output impedance atterminal 67 as the preferred embodiment shown in FIG. 2.

FIG. 4 is also another alternate embodiment of op amp input stage 12.MOSFET 30 is coupled to resistor 208 and MOSFET 32 is coupled toresistor 210. The second terminals of resistors 208 and 210 are coupledto operating potential V_(CC). The differential pair of MOSFETS 30 and32 that receive the input signal V_(IN) provide outputs from the drainterminals of MOSFETS 30 and 32. The output from the drain of MOSFET 30is coupled to the emitter of PNP transistor 212. The output from thedrain of MOSFET 32 is coupled to the emitter of PNP transistor 214.Common bases of transistors 212 and 214 are coupled to the collector oftransistor 212. The first terminal of current sink 216 is coupled to thecollector of transistor 212. The collector of transistor 214 is coupledto the output terminal 67 for providing the signal STAGE-1 OUTPUT. Thefirst terminal of current sink 218 is coupled to terminal 67. The secondterminals of current sink 216 and 218 are coupled to ground reference.Again, the alternate embodiment shown in FIG. 4 does not match sourceand sink current capabilities or provide as high an output impedance atterminal 67 as the preferred embodiment shown in FIG. 2.

FIG. 5 shows a schematic diagram of sink control circuit 14 suitable foruse in low voltage operational amplifier 10 of FIG. 1. Common bases ofNPN transistors 74, 76, 78, and 80, are coupled to receive as input tosink control circuit 14 the output from the source of MOSFET 13, asshown in FIG. 1. The emitter of transistor 74 is coupled to the firstterminal of resistor 82, selected at approximately three ohms in thepreferred embodiment. The emitter of transistor 76 is coupled to thefirst terminal of resistor 84, selected at approximately one and onehalf kilohms. The emitter of transistor 78 is coupled to the firstterminal of resistor 86, selected at approximately one and one halfkilohms. The emitter of transistor 80 is coupled to the first terminalof resistor 88, selected at approximately one and one half kilohms. Thesecond terminals for resistors 82, 84, 86, and 88, are coupled to groundreference.

Common bases of NPN transistors 90 and 92 in FIG. 5 are coupled to thefirst terminal of resistor 94, selected at approximately twenty fivekilohms. The emitter of transistor 90 is coupled to the collector oftransistor 74. Common emitters of transistors 92 and 96 are coupled tothe collector of transistor 76. The collector of transistor 92 iscoupled to the emitter of PNP transistor 100 and to the first terminalof resistor 98, selected at approximately four kilohms. The collector ofNPN transistor 96 is coupled to the emitter of PNP transistor 102 and tothe first terminal of resistor 104, selected at approximately fourkilohms. Common bases of transistors 100 and 102 are coupled to thecollector of transistor 100 and to the collector of transistor 78. Thecollector of transistor 102 couples to the collector of transistor 80and to the base of PNP transistor 106. The first terminal of capacitor108 selected at approximately five picofarads capacitance, couples tothe base of transistor 106. The second terminal of capacitor 108 iscoupled to ground reference. The collector of transistor 106 is coupledto terminal 107, which provides the signal SINK-1 PASS THROUGH. Theemitter of transistor 106 is coupled to the first terminal of resistor110, selected at approximately twenty five kilohms, and to the firstterminal of resistor 112, selected at approximately one kilohm. Thesecond terminal of resistor 110 is coupled to the base of transistor 96.The second terminal of resistors 94, 98, 104, and 112, and the collectorof transistor 90 are coupled to the operating potential V_(CC).

The function of sink control circuit 14 in FIG. 5 is to supply theproper base drive current required by output transistor 18, shown inFIG. 1, for sinking a current such as I_(out) at the output of lowvoltage operational amplifier 10. The emitter geometry of transistor 18in FIG. 1 is sized at N_(T) times the emitter geometry of transistor 74in FIG. 5. For this preferred embodiment, the N_(T) transistormultiplier for ratioing is approximately twenty five. Thus, outputtransistor 18 has a collector current N_(T) times greater than thecollector current of transistor 74. Transistor 90 is sized with the sameor similar emitter geometry as transistor 74, and therefore conducts thesame or similar collector current I_(out) /N_(T). The base current oftransistor 90 is I_(out) /(N_(T) ·B), where B is the transistor currentgain defined as the ratio of transistor collector current divided bytransistor base current. Transistors 92 and 96 form a differential unitygain amplifier with the base of transistor 92 sensing the voltage dropresulting from the I_(out) /(N_(T) ·B) current in resistor 94.

Thus, transistor 90 and resistor 94 have converted a proportionatelysmaller current than the I_(out) found in transistor 18 into a voltageacross resistor 94 which becomes one input to the differential unitygain amplifier. The voltage at the base of transistor 92 is the currentthrough resistor 94 multiplied by the resistance R₉₄ of resistor 94, fora voltage of (I_(out) ·R₉₄)/(N_(T) ·B). Both inputs to the differentialunity gain amplifier have matching voltage potentials. The other inputto the differential unity gain amplifier is applied at the base oftransistor 96. The voltage at the base of transistor 96 results fromcurrent I_(C) flowing through resistor 112, having a resistance R₁₁₂.With both inputs to the differential unity gain amplifier havingmatching voltage potentials, the result is (I_(C) ·R₁₁₂)=(I_(out)·R₉₄)/(N_(T) ·B). Solving for the current I_(C) results in (I_(out)·N_(R))/(N_(T) ·B) where N_(R) is the ratio of resistance values forresistor 94 and resistor 112, a value of R₉₄ /R₁₁₂. The current I_(C)through resistor 112 essentially becomes the emitter-to-collectorcurrent of transistor 106. By selecting the value N_(R) to match N_(T),the current I_(C) has the value of I_(out) /B. Thus, by matching theratio of two transistors, transistor 18 and transistor 74, to the ratioof two resistors, namely resistor 94 and resistor 112, the currentI_(out) /B through transistor 106 supplies the base current to sinktransistor 18. With a base current of I_(out) /B in transistor 18 asshown in FIG. 1, collector current for transistor 18 is I_(out). Thefunction of sink control circuit 14 in FIG. 5 is to supply the properbase drive current required by output transistor 18, shown in FIG. 1,for sinking current I_(out) at the output of low voltage operationalamplifier 10.

Thus, sink control circuit 14 accomplishes three transformation steps.The first step involves providing transistor emitter geometry ratios fortransistor 18 and transistor 74 to generate a current of I_(out) /(N_(T)·B) in the base of transistor 90. In step two, sink control circuit 14generates a voltage at inputs to differential unity gain amplifierdependent on the generated I_(out) /(N_(T) ·B) current in resistor 94.The final step involves resistor ratioing such that transistor 106 insink control circuit 14 generates a collector current I_(out) /B intransistor 106 for supplying base drive current to output transistor 18in low voltage operational amplifier 10. Such a base drive current fortransistor 18 shown in FIG. 1 is dependent on both transistor andresistor ratioing and the voltage developed by differential unity gainamplifier found in sink control circuit 14 shown in FIG. 5. For thispreferred embodiment, N_(T) transistor ratioing is approximately twentyfive and N_(R) resistor ratioing is approximately twenty five.

In low voltage operational amplifier 10 in FIG. 1, amplification of theinput signal V_(IN) provides the signal STAGE-1 OUTPUT at terminal 67 asthe op amp input stage 12 output, which MOSFET 13 passes directly to thebase of transistor 18, causing a base-to-emitter voltage (V_(be))change. The V_(be) change causes transistor 18, sinking a currentI_(out), to modify the current and sink (I_(out) +ΔI_(out)). Sinkcontrol circuit 14 responds to a ΔV_(be) at the base of transistor 18and generates the additional base current for transistor 18 inaccounting for ΔI_(out) collector current changes in sink transistor 18.Sink control circuit 14 supplies the base drive current throughtransistor 106 as required by output sink transistor 18 shown in FIG. 1as low voltage operational amplifier 10 responds to changes to the inputsignal VIN.

Source control circuit 22 as shown in FIG. 1 is shown in FIG. 6 as apreferred embodiment. Common bases of PNP transistors 114, 116, 118, and120, are coupled to terminal 147 that provides the signal, SOURCE-1 PASSTHROUGH. The emitter of transistor 114 is coupled to the first terminalof resistor 122, selected at approximately ten ohms. The emitter oftransistor 116 is coupled to the first terminal of resistor 124,selected at approximately four kilohms. The emitter of transistor 118 iscoupled to the first terminal of resistor 126, selected at approximatelyone kilohm. The emitter of transistor 120 is coupled to the firstterminal of resistor 128, selected at approximately one kilohm. Thesecond terminals for resistors 122, 124, 126, and 128 are coupled tooperating potential V_(CC).

Common bases of PNP transistors 130 and 132 are coupled to the firstterminal of resistor 134, selected at approximately twenty five kilohms.The emitter of transistor 130 is coupled to the collector of transistor114. Common emitters of transistors 132 and 136 are coupled to thecollector of transistor 116. The collector of transistor 132 is coupledto the emitter of transistor 140 and to the first terminal of resistor138, selected at approximately four kilohms. The collector of PNPtransistor 136 is coupled to the emitter of transistor 142 and to thefirst terminal of resistor 144, selected at approximately four kilohms.Common bases of NPN transistors 140 and 142 are coupled to the collectorof transistor 140 and to the collector of transistor 118. The collectorof transistor 142 couples to the collector of transistor 120 and to thebase of NPN transistor 146. Capacitor 148, selected at a capacitance ofapproximately ten picofarads, has the first terminal coupled to the baseof transistor 146. The second terminal of capacitor 148 is coupled toground reference. The collector of transistor 146 is coupled to terminal147 providing the signal, SOURCE-1 PASS THROUGH. The emitter oftransistor 146 is coupled to the first terminal of resistor 150,selected at twenty five kilohms, and to the first terminal of resistor152, selected at approximately five hundred ohms. The second terminal ofresistor 150 is coupled to the base of transistor 136. The secondterminal of resistors 134, 138, 144, and 152, and the collector oftransistor 130 are coupled to ground reference.

The function of source control circuit 22 in FIG. 6 is to supply theproper base drive current required by output transistor 24, shown inFIG. 1, for sourcing a current such as I_(out) at the output of lowvoltage operational amplifier 10. The emitter geometry of transistor 24in FIG. 1 is sized at N_(t) times the emitter geometry of transistor 114in FIG. 6. For this preferred embodiment, the N_(t) transistor ratioingmultiplier is approximately fifty. Thus, output transistor 24 has acollector current N_(t) times greater than the collector current oftransistor 114. Transistor 130 is sized with the same or similar emittergeometry as transistor 114, and therefore conducts the same or similarcollector current I_(out) /N_(t). The base current of transistor 130 isI_(out) /(N_(t) ·B), where B is the transistor current gain defined asthe ratio of transistor collector current divided by transistor basecurrent. Transistors 132 and 136 form a differential unity gainamplifier with the base of transistor 132 sensing the voltage dropresulting from the I_(out) /(N_(t) ·B) current in resistor 134.

Thus, transistor 130 and resistor 134 have converted a proportionatelysmaller current than the I_(out) found in transistor 24 into a voltageacross resistor 134 which becomes one input to the differential unitygain amplifier. Therefore, the voltage at the base of transistor 132 isthe current through resistor 134 multiplied by the resistance R₁₃₄ ofresistor 134, for a voltage of (I_(out) ·R₁₃₄)/(N_(t) ·B). Both inputsto the differential unity gain amplifier have matching voltagepotentials. The other input to the differential unity gain amplifier isapplied at the base of transistor 136. The voltage at the base oftransistor 136 results from current I_(c) flowing through resistor 152,having a resistance R₁₅₂. With both inputs to the differential unitygain amplifier have matching voltage potentials, the result is (I_(c)·R₁₅₂)=(I_(out) ·R₁₃₄)/(N_(t) ·B). Solving for the current I_(c) resultsin (I_(out) ·N_(r))/(N_(t) ·B) where N_(r) is the ratio of resistancevalues for resistor 134 and resistor 152, a value of R₁₃₄ /R₁₅₂. Thecurrent I_(c) through resistor 152 essentially becomes thecollector-to-emitter current of transistor 146. By selecting the valueN_(r) to match N_(t), the current I_(c) has the value of I_(out) /B.Thus, by matching the ratio of two transistors, transistor 24 andtransistor 114, to the ratio of two resistors, namely resistor 134 andresistor 152, the current I_(out) /B through transistor 146 supplies thebase current to source transistor 24. With a base current of I_(out) /Bin transistor 24 as shown in FIG. 1, collector current for transistor 24is I_(out). The function of source control circuit 22 in FIG. 6 is tosupply the proper base drive current through transistor 146 as requiredby output transistor 24, shown in FIG. 1, for sourcing current I_(out)at the output of low voltage operational amplifier 10.

Thus, source control circuit 22 accomplishes three transformation steps.The first step involves providing transistor emitter geometry ratios fortransistor 24 and transistor 114 to generate a current of I_(out)/(N_(t) ·B) in the base of transistor 130. In step two, source controlcircuit 22 generates a voltage at inputs to differential unity gainamplifier dependent on the generated I_(out) /(N_(t) ·B) current inresistor 134. The final step involves resistor ratioing of resistors 152and 134 such that transistor 146 in source control circuit 22 generatesa collector current I_(out) /B for supplying base drive current tooutput transistor 24 in low voltage operational amplifier 10. Such abase drive current for transistor 24 shown in FIG. 1 is dependent onboth transistor and resistor ratioing and the voltage developed bydifferential unity gain amplifier found in source control circuit 22shown in FIG. 6. For this preferred embodiment, N_(t) transistorratioing is approximately fifty and N_(r) resistor ratioing isapproximately fifty.

In low voltage operational amplifier 10 in FIG. 1, amplification of theinput signal V_(IN) provides the signal STAGE-1 OUTPUT as the op ampinput stage 12 output, which MOSFET 13 passes directly to the base oftransistor 18, causing a base-to-emitter voltage (V_(be)) change.Translinear loop 16 passes the same magnitude V_(be) voltage changefound at the base of transistor 18 onto the base of transistor 24.However, the V_(be) voltage change has the opposite sign, i.e., ifV_(be) for transistor 18 is increasing the V_(be) for transistor 24 isdecreasing. The V_(be) change causes transistor 24, sourcing a currentI_(out), to modify the current and source (I_(out) -ΔI_(out)). Sourcecontrol circuit 22 supplies the base drive current required by outputsource transistor 24 shown in FIG. 1 as low voltage operationalamplifier 10 responds to changes to the input signal VIN.

FIG. 7 shows an embodiment of simplified translinear loop 16. The baseof NPN transistor 230 is coupled to terminal 107. The common collectorsof NPN transistors 230 and 232 are coupled to the common bases of NPNtransistors 232 and 234. The common emitters of transistors 230, 232,and 234 are coupled to ground reference. Current source 236 is coupledto the collector of transistor 232. The second terminal of currentsource 236 is coupled to operating potential V_(CC). The base andcollector of PNP transistor 238 are coupled to the collector oftransistor 234. The emitter of transistor 238 is coupled to operatingpotential V_(CC). The base and collector of PNP transistor 238 arecoupled to output terminal 147. Terminal 147 is coupled to the base ofsource transistor 24 of output driver stage 29 (see FIG. 1).

Still referring to FIG. 7, as an example, the simplified embodiment oftranslinear loop 16 receives a positive voltage change at terminal 107which modifies the base-to-emitter voltage V_(be) of transistor 230. Thesame +ΔV_(be) that causes transistor 18 in output driver stage 29 (seeFIG. 1) to increase in conductivity also causes transistor 230 toincrease conductivity and shunt current from diode connected transistor232. Thus, current source 236 supplies current that transistor 230proportionately steers into the collector terminal of transistor 230 ordiverts into transistor 232 as determined by the ΔV_(be) of transistor230 from the received signal at terminal 107. Transistor 234 forms acurrent mirror transistor with transistor 232. The +ΔV_(be) attransistor 230 causes a decreasing current conducted by transistor 232,and the current mirror causes a decreasing current conducted bytransistor 234. Decreased current in transistor 234 means decreasedcurrent in diode connected transistor 238, causing a decreased V_(be) intransistor 238. The same decreasing V_(be) seen at the base oftransistor 238 is seen at the base of output source transistor 24 inoutput driver stage 29 (see FIG. 1). Therefore, an increasing +ΔV_(be)for a higher conductivity in output sink transistor 18 (see FIG. 1) istranslated into an equivalent decreasing -ΔV_(be) for a lowerconductivity in output source transistor 24 (see FIG. 1) by translinearloop 16.

When the simplified embodiment of translinear loop 16 shown in FIG. 7receives a negative voltage change at terminal 107, the base-to-emittervoltage V_(be) of transistor 230 is modified. The same -ΔV_(be) thatcauses transistor 18 in output driver stage 29 (see FIG. 1) to decreaseconductivity also causes transistor 230 to decrease conductivity, whichincreases current to diode connected transistor 232. Thus, currentsource 236 supplies current that transistor 230 proportionately steersinto the collector terminal of transistor 230 or diverts into transistor232 as determined by the V_(be) change of transistor 230 caused by thereceived signal at terminal 107. Transistor 234 forms a current mirrortransistor with transistor 232. The -ΔV_(be) at transistor 230 thereforecauses an increase in current conducted by transistor 234. Increasedcurrent in transistor 234 means increased current in diode connectedtransistor 238, causing an increased V_(be) in transistor 238. The sameincreasing V_(be) seen at the base of transistor 238 is seen at the baseof output source transistor 24 in output driver stage 29 (see FIG. 1).Therefore, a decreasing V_(be) for a decreasing conductivity in outputsink transistor 18 (see FIG. 1) is translated into an equivalent+ΔV_(be) for an increasing conductivity in output source transistor 24(see FIG. 1) by translinear loop 16.

With reference to FIG. 7, quiescent currents for low voltage translinearloop 16 rely on relationships for sizing the geometry of a transistor.The emitter area of transistor 18 (see FIG. 1) is sized at N_(n) timesthe emitter area of transistor 230. The emitter area of transistor 24(see FIG. 1) is sized at N_(p) times the emitter area of transistor 238.Also, the current mirror transistors are sized such that emittergeometry of transistor 234 is M_(n) times the emitter geometry oftransistor 232. Since the area of an emitter determines the currentcapacity for a transistor, the current 2I out of current source 236 andthe selection of three variables N_(n), N_(p), and M_(n) set the othercurrents in low voltage translinear loop 16. Thus, the quiescent currentI_(Q) in sink transistor 18 (see FIG. 1) is set by I_(Q) =(N_(n) ·I) andthe quiescent current I_(Q) in source transistor 24 (see FIG. 1) is setby I_(Q) =(M_(n) ·N_(p) ·I). Adding resistors in the coupling path ofemitter terminal to ground reference for transistors 230, 232, and 234or adding a resistor in the coupling path of emitter terminal fortransistor 238 to operating potential V_(CC) causes emitter degenerationand allows multiplier factors N_(n), N_(p), and M_(n) to change.

FIG. 8 shows the preferred embodiment of translinear loop 16 asmentioned in FIG. 1. Common bases of PNP transistors 154 and 156 arecoupled to the collector of transistor 154 and to the first terminal ofcurrent sink 158, sinking approximately ten microamps of current. Thecollector of transistor 156 is coupled to the base of NPN transistor 160and to the first terminal of resistor 162, selected at approximatelythirty-three kilohms. The second terminal of resistor 162 couples to thebase and collector of NPN transistor 164. The emitter of transistor 160couples to the collector of NPN transistor 166. The base of transistor166 couples to terminal 107 for receiving the signal SINK-1 PASSTHROUGH. The emitter of transistor 160 couples to the collector of PNPtransistor 168. The emitter of transistor 160 couples to the commonbases of NPN transistors 170 and 172. The emitter of transistor 160couples to the collector of transistor 170 and to the first terminal ofcurrent source 174, sourcing approximately one hundred and seventy fivemicroamps. The emitter of transistor 166 couples to the first terminalof resistor 176, selected at approximately fifty ohms. The emitter oftransistor 170 is coupled to the first terminal of resistor 178,selected at approximately one hundred ohms. The emitter of transistor172 is coupled to the first terminal of resistor 180, selected atapproximately twenty-five ohms. The emitter of transistor 168 is coupledto resistor 182, selected at approximately three-hundred ohms. Thecommon collectors of transistors 172 and 184 are coupled to the base ofPNP transistor 184 and coupled to terminal 147 for providing the signalSOURCE-1 PASS THROUGH. The emitter of transistor 184 is coupled to thefirst terminal of resistor 186, selected at approximately four-hundredohms. The emitters of transistors 154 and 156 are coupled to operatingpotential V_(CC). The collector of transistor 160 is coupled tooperating potential V_(CC). The second terminals of resistor 182 and 186and the second terminal of current source 174 are coupled to operatingpotential V_(CC). The second terminals of resistors 176, 178, and 180are coupled to ground reference. The emitter of transistor 164 and thesecond terminal of current sink 158, are coupled to ground reference.

Translinear loop in FIG. 8 provides a fast output stage with highfrequency response characteristics. In a manner similar to the alreadymentioned simplified embodiment of translinear loop 16, an increasingvoltage signal SINK-1 PASS THROUGH at terminal 107 causes transistor 166to shunt current away from diode connected transistor 170. Less currentin transistor 170 also means less current in the current mirror device,transistor 172. The decrease in transistor 172 current means lowercurrent in diode connected transistor 184, causing a lower V_(be)voltage in transistor 184. The lower base-to-emitter voltage fortransistor 184 is also seen as the V_(be) for transistor 24 shown inFIG. 1. Thus, an AC signal modulating the base voltage of transistor 18to a more positive potential causes transistor 18 to be more conductive,but translinear loop 16 causes transistor 24 to be less conductive.Translinear loop 16 transposes AC signals from the base of transistor 18to the base of transistor 24 without providing signal voltage gain. Onlyop amp input stage 12 and output transistors 18 and 24, provide signalgain. A +ΔV_(be) across sink transistor 18 (see FIG. 1) due to signalSINK-1 PASS THROUGH at terminal 107 is translated to a matching -ΔV_(be)across source transistor 24 (see FIG. 1) by translinear loop 16.

In a manner similar to the already mentioned simplified embodiment oftranslinear loop 16, a decreasing voltage signal SINK-1 PASS THROUGH atterminal 107 causes transistor 166 to steer current to diode connectedtransistor 170. More current in transistor 170 also means more currentin the current mirror device, transistor 172. The increase in transistor172 current means higher current in diode connected transistor 184,causing a higher V_(be) in transistor 184. The increased base-to-emittervoltage for transistor 184 is also seen as the V_(be) for transistor 24shown in FIG. 1. Thus, an AC signal modulating the base voltage oftransistor 18 to a lower voltage potential causes transistor 18 to beless conductive, but translinear loop 16 causes transistor 24 to be moreconductive. A -ΔV_(be) across sink transistor 18 (see FIG. 1) due tosignal SINK-1 PASS THROUGH at terminal 107 is translated to a matching+ΔV_(be) across source transistor 24 (see FIG. 1) by translinear loop16. Low voltage translinear loop 16 provides a low impedance path tooutput devices, thus ensuring no voltage gain to the base of sourcingtransistor 24.

Sink control circuit 14 and source control circuit 22 in FIG. 1 provideimportant Direct Current (DC) generating functions in providing basecurrent drive for the output transistors 18 and 24 in output driverstage 29. However, low voltage operational amplifier 10 frequencyperformance does not depend on sink control circuit 14 or source controlcircuit 22. Low voltage operational amplifier 10 frequency performancedepends on the AC signal path from V_(IN) of op amp input stage 12 tothe STAGE-1 OUTPUT, through source follower MOSFET 13, directly to thebase of output current sinking transistor 18. The AC signal path fromthe current sink side to the current source side follows the base ofoutput current sinking transistor 18, through translinear loop 16, tothe base of output current sourcing transistor 24. Thus, the AC signalpath bypasses the circuitry in sink control circuit 14 and sourcecontrol circuit 22, allowing a higher frequency performance in lowvoltage operational amplifier 10. The bandwidth of low voltageoperational amplifier 10 is five megahertz. Bias circuit 23 is comprisedof sink control circuit 14, source control circuit 22, and translinearloop 16. A first bias output is generated at terminal 107 in accordancewith the signal transferred across the source follower and the currentgenerated by sink control circuit 14. A second bias output is generatedat terminal 147 in accordance with the signal transferred by translinearloop 16 and the current generated by source control circuit 22.

By now it should be appreciated that low voltage operational amplifier10 in FIG. 1 operates in a voltage range of eight volts to one volt overa temperature range of 0° to 70° centigrade. Op amp input stage 12 usesN-channel depletion-mode MOSFETs 30 and 32 (see FIG. 2) to provideamplification of the input V_(IN) and maintain constanttransconductance. Source follower MOSFET 13 (see FIG. 1) provides unitygain in transferring the AC signal, STAGE-1 OUTPUT, to the base ofcurrent sinking transistor 18. A separate DC loop through sink controlcircuit 14 and source control circuit 22, creates the bias for the basedrive current in transistors 18 and 24. Dependent on the input signalthe AC signal path on SINK PASS THROUGH signal controls op amp outputsink transistor to sink current or through translinear loop 16 causesSOURCE PASS THROUGH signal controlling op amp output source transistorto source current. An output stage provides approximately fiftymilliamps of sink and source current.

While the invention has been described in the context of a preferredembodiment, it will be apparent to those skilled in the art that thepresent invention may be modified in numerous ways and may assume manyembodiments other than that specifically set out and described above.Accordingly, it is intended by the appended claims to cover allmodifications of the invention which fall within the true spirit andscope of the invention.

What is claimed is:
 1. A low voltage operational amplifier comprising:aninput stage having an input coupled for receiving an input signal andhaving an output, the input stage including:a first depletion-mode metaloxide semiconductor field effect transistor (MOSFET) having a sourceterminal, a drain terminal, and a gate terminal; and a seconddepletion-mode MOSFET having a source terminal, a drain terminal, and agate terminal, wherein the source terminals of the first and seconddepletion-mode MOSFETs are commonly coupled to form a differential pairand the gate terminals serve as differential inputs of the differentialpair; a source follower having a gate terminal, a drain terminal, and asource terminal, wherein the gate terminal of the source follower iscoupled to the input stage; a bias circuit having an input coupled tothe source terminal of the source follower, wherein the bias circuitprovides a first bias output signal and a second bias output signal; andan output driver stage coupled to the bias circuit and to the sourceterminal of the source follower, wherein the output driver stagereceives the first bias output signal and the second bias output signaland provides an output signal.
 2. A low voltage operational amplifier asclaimed in claim 1, wherein the source follower is a depletion-modeMOSFET.
 3. A low voltage operational amplifier as claimed in claim 2,wherein the source follower is processed in a P-type well.
 4. A lowvoltage operation amplifier as claimed in claim 1, further comprising afrequency pole splitting circuit coupled between the output driver stageand the op amp input stage output.
 5. A low voltage operationalamplifier as claimed in claim 1, further comprising a current sinkcoupled between the source terminal of the source follower and groundreference.
 6. A low voltage operational amplifier as claimed in claim 1,wherein the low voltage operational amplifier operates at a voltage downto one volt.
 7. A low voltage operational amplifier as claimed in claim1, wherein the low voltage operational amplifier operates over atemperature range of about zero degrees Celsius to about 70 degreesCelsius.
 8. A low voltage operational amplifier as claimed in claim 1,wherein the bias circuit comprises:a source control circuit providing asource pass through; a sink control circuit coupled to the sourcefollower, the sink control circuit for providing a sink pass through;and a translinear loop coupled to the sink control circuit and to thesource control circuit, the translinear loop for receiving the sink passthrough and for producing the second bias output therefrom.
 9. A lowvoltage operational amplifier as claimed in claim 1, wherein the op ampinput stage comprises N-channel depletion-mode Metal Oxide SemiconductorField Effect Transistors (MOSFETs) to provide amplification and toexhibit minimal transconductance changes.
 10. A method for amplifying adifferential input signal in a low voltage operational amplifier, themethod comprising the steps of:receiving the differential input signal;amplifying the differential input signal using N-channel depletion-modemetal oxide semiconductor field effect transistors (MOSFETs) to generatean output signal of the input stage; transferring the output signal ofthe input stage to a translinear loop and an output driver stage usingan N-channel depletion-mode MOSFET; transferring an output signal of theN-channel depletion-mode MOSFET source follower to the output driverstage; and generating an output signal from an output driver stage. 11.A method as claimed in claim 10, wherein the step of providing a firstbias output and a second bias output from the op amp input stage outputcomprises the steps of:generating a source pass through from a sourcecontrol circuit; generating a sink pass through from a sink controlcircuit; and steering the sink pass through and the source pass throughin response to the differential input to produce the first bias outputand the second bias output.
 12. A method for transferring an alternatingcurrent (AC) signal from an input of an operational amplifier (op amp)input stage to a bias circuit, the method comprising the stepsof:receiving the AC signal at the input of the op amp input stage; andtransferring an output signal of the op amp input stage to the biascircuit using a depletion-mode metal oxide semiconductor field effecttransistor (MOSFET) source follower.
 13. A method as claimed in claim12, wherein the step of transferring the output signal of the op ampinput stage to the bias circuit includes transferring substantially allof the output signal of the op amp input stage to the bias circuit. 14.A method as claimed in claim 12, wherein the step of transferring theoutput signal of the op amp input stage further includes using anN-channel depletion-mode MOSFET configured as a source follower.
 15. Amethod as claimed in claim 14, wherein the step of transferring theoutput of an op amp input stage to the bias circuit includes operatingthe op amp input stage at a voltage down to one volt.
 16. A low voltageoperational amplifier (op amp) comprising:an op amp input stage forreceiving an input and providing an amplified output signal of the opamp input stage, wherein the input is received by a differential pair ofmetal oxide semiconductor field effect transistors (MOSFETs); adepletion-mode source follower having a gate terminal, a drain terminal,and a source terminal, wherein the gate terminal of the source followeris coupled to the input stage and receives the amplified output signaland provides an output; a bias circuit coupled to the output of thedepletion-mode source follower, the bias circuit for receiving theamplified output signal of the op amp input stage and for providing afirst bias output and a second bias output; and an output driver stagecoupled to the bias circuit and to the op amp input stage output, theoutput driver stage for receiving the first bias output and the secondbias output and for providing an output driver stage output.
 17. A lowvoltage operation amplifier as claimed in claim 16, further comprising afrequency pole splitting circuit coupled between the output driver stageand the op amp input stage output.
 18. A low voltage operationalamplifier as claimed in claim 16, wherein the low voltage operationalamplifier operates in a voltage range of about one volt to about eightvolts.
 19. A low voltage operational amplifier as claimed in claim 16,wherein the bias circuit comprises:a source control circuit providing asource pass through; a sink control circuit coupled to the op amp inputstage, the sink control circuit for providing a sink pass through; and atranslinear loop coupled to the sink control circuit and to the sourcecontrol circuit, the translinear loop for receiving the sink passthrough and for producing the second bias output therefrom.
 20. A lowvoltage operational amplifier as claimed in claim 16, wherein the op ampinput stage and the output driver stage provides signal amplification.